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T H A T Corporation
FEATURES
* * * * * * * High-Performance Voltage Controlled Amplifier High-Performance RMS-Level Detector Three General-Purpose Opamps Wide Dynamic Range: >115 dB Low THD: <0.03% Low Cost: $4.39 (`000s) DIP & Surface-Mount Packages * * * * * * * *
THAT Analog Engine(R) IC Dynamics Processor
THAT 4301, 4301A APPLICATIONS
Compressors Limiters Gates Expanders De-Essers Duckers Noise Reduction Systems Wide-Range Level Meters
Description
THAT 4301 Dynamics Processor, dubbed "THAT Analog Engine," combines in a single IC all the active circuitry needed to construct a wide range of dynamics processors. The 4301 includes a high-performance, exponentially-controlled VCA, a log-responding RMS-level sensor and three general- purpose opamps. The VCA provides two opposing-polarity, voltage-sensitive control ports. Dynamic range exceeds 115 dB, and THD is typically 0.003% at 0 dB gain. In the 4301A, the VCA is selected for low THD at extremely high levels. The RMS detector provides accurate rms-to-dc conversion over an 80 dB dynamic range for signals with crest factors up to 10. One opamp is dedicated as a current-to-voltage converter for the VCA, while the other two may be used for the signal path or control voltage processing. The combination of exponential VCA gain control and logarithmic detector response -- "decibel-linear" response -- simplifies the mathematics of designing the control paths of dynamics processors. This makes it easy to design audio compressors, limiters, gates, expanders, de-essers, duckers, noise reduction systems and the like. The high level of integration ensures excellent temperature tracking between the VCA and the detector, while minimizing the external parts count.
18
11
17
14
13
12
19
+
OA1
VCC IN EC-
SYM VCA OUT EC+
+
OA3
15
20
THAT4301
1
IN IT
2
RMS
+
OUT CT
5 4
16
OA2
GND VEE
9 10 8 6
-
7
Figure 1. Block Diagram (pin numbers are for DIP only)
THAT Corporation; 45 Sumner Street; Milford, Massachusetts 01757-1656; USA Tel: +1 (508) 478-9200; Fax: +1 (508) 478-0990; Web: www.thatcorp.com
Page 2
Rev. 04/10/02
SPECIFICATIONS 1 , 2 Absolute Maximum Ratings (T A = 25C)
Positive Supply Voltage (VCC) Negative Supply Voltage (VEE) Supply Current (ICC) +18 V -18 V 20 mA Power Dissipation (PD) (TA = 75C) Operating Temperature Range (TOP) Storage Temperature Range (TST) 700 mW 0 to +70C -40 to +125C
Overall Electrical Characteristics
Parameter Positive Supply Voltage Negative Supply Voltage Positive Supply Current Negative Supply Current Thermal Resistance Symbol VCC VEE ICC IEE qJ-C SO-Package Conditions Min +7 -7 -- -- -- Typ -- -- 12 -12 140 Max +15 -15 18 -18 -- Units V V mA mA C/W
VCA Electrical Characteristics 3
4301 Parameter Input Bias Current Input Offset Voltage Input Signal Current Gain at 0V Control Gain-Control Constant EC+/Gain (dB) EC-/Gain (dB) Gain-Control TempCo Gain-Control Linearity Off Isolation Output Offset Voltage Change DEC / DTCHIP Symbol IB(VCA) VOFF(VCA In) IIN(VCA) or IOUT(VCA) G0 EC+ = EC- = 0.000V TA = 25C (TCHIP @ 55C) -60 dB < gain < +40dB EC+ & SYM ECRef TCHIP= 27C -60 to +40 dB gain 6.4 -6.4 -- -- 6.5 -6.5 +0.33 0.5 115 6.6 -6.6 -- 2 -- 6.4 -6.4 -- -- 110 6.5 -6.5 +0.33 0.5 115 6.6 -6.6 -- 2 -- mV/dB mV/dB %/C % dB Conditions No Signal No Signal Min -- -- -- -0.4 Typ 30 4 175 0.0 Max 400 15 750 +0.4 Min -- -- -- -0.4 4301A Typ 30 4 175 0.0 Max 400 15 750 +0.4 Units pA mV mArms dB
EC+=SYM=-375mV, EC-=+375mV 110 DVOFF(OUT) Rout = 20kW 0 dB gain +15 dB gain +30 dB gain -- -- -- -- 20 Hz-20 kHz Rout = 20kW 0 dB gain +15 dB gain -- --
1 2 5 20
3 10 25 --
-- -- -- --
1 2 5 20
3 10 25 --
mV mV mV mA
Gain Cell Idling Current Output Noise
IIDLE en(OUT)
-96 -85
-94 -83
-- --
-96 -85
-94 -83
dBV dBV
Total Harmonic Distortion
THD
VIN= 0 dBV, 1 kHz 0 dB gain -- 0.003 0.007 -- 0.003 0.007 %
1. All specifications subject to change without notice. 2. Unless otherwise noted, TA=25C, VCC = +15V, VEE= -15V; VCASYM adjusted for min THD @ 1 V, 1 kHz, 0 dB gain. 3. Test circuit is the VCA section only from Figure 2.
THAT Corporation; 45 Sumner Street; Milford, Massachusetts 01757-1656; USA Tel: +1 (508) 478-9200; Fax: +1 (508) 478-0990; Web: www.thatcorp.com
THAT 4301 Dynamics Processor IC
Page 3
SPECIFICATIONS 1 , 2 (Cont'd.) VCA Electrical Characteristics 3 (Cont'd.)
Parameter Symbol Conditions VIN = +10 dBV, 1 kHz 0 dB gain -15 dB gain VOUT= +10 dBV, 1 kHz +15 dB gain VIN = +19.5 dBV, 1 kHz 0 dB gain Symmetry Control Voltage VSYM minimum THD -- -- -2.5 0.035 0.09 -- 0 -- +2.5 -- -- -2.5 0.035 0.09 0.05 0 0.09 +2.5 % % mV -- -- 0.03 0.07 -- -- 0.03 0.07 % % 0.035 0.09 0.035 0.09 Min 4301 Typ Max Min 4301A Typ Max Units
Total Harmonic Distortion (cont'd.) THD
RMS Detector Electrical Characteristics 4
Parameter Input Bias Current Input Offset Voltage Input Signal Current Input Current for 0 V Output Output Scale Factor Symbol IB (RMS) VOFF(RMS In) IIN(RMS) Iin0 EO / 20log(Iin/Iin0) IT= 7.5mA 31.6nA< IIN< 1mA TA= 25C (TCHIP 55C) Scale Factor Match (RMS to VCA) -20 dB < VCA Gain < +20 dB 1mA( 0.026)CT IT
0.33 100
4. Except as noted, test circuit is the RMS-Detector section only from Figure 2. THAT Corporation; 45 Sumner Street; Milford, Massachusetts 01757-1656; USA Tel: +1 (508) 478-9200; Fax: +1 (508) 478-0990; Web: www.thatcorp.com
Page 4
Rev. 04/10/02
Specifications 1 , 2 (Cont'd) Opamp Electrical Characteristics 5
OA1 Parameter Input Offset Voltage Input Bias Current Input Offset Current Input Voltage Range Symbol VOS IB IOS IVR RS<10k VS=7V to 15V (@50kHz) RL=10k RL=2k Output Voltage Swing VO@RL=5kW VO@RL=2kW Short Circuit Output Current Slew Rate Total Harmonic Distortion SR THD -- -- -- Conditions Min -- -- -- Typ Max 0.5 150 15 6 500 50 Min -- -- -- OA2 Typ Max 0.5 150 15 6 500 50 Min -- -- OA3 Typ Max 0.5 150 N/A N/A N/A -- -- -- -- -- -- -- -- -- -- -- -- 100 5 125 120 14 13 12 2 -- -- -- -- -- -- -- -- V V mA V/ms MHz 6 500 Units mV nA nA V
-- 13.5 -- -- -- -- -- 100 100 5 115 N/A 13 N/A 4 2 -- -- -- -- -- -- --
-- 13.5 -- -- -- -- -- 100 100 5 110 N/A -- 13 N/A -- -- 4 2 -- -- -- --
Common Mode Rej. Ratio CMRR Power Supply Rej. Ratio Gain Bandwidth Product Open Loop Gain PSRR GBW AVO
1kHz, AV=1, RL=10kW -- 0.0007 0.003 -- 0.0007 0.003 -- 0.0007 0.003 % 1kHz, AV=-1, RL= 2kW N/A -- -- 6.5 0.3 10 -- -- -- N/A 7.5 0.3 12 -- -- 0.0007 0.003 % -- -- 7.5 0.3 12 -- nV pA Hz Hz
Input Noise Voltage Density Input Noise Current Density
en in
fO=1kHz fO=1kHz
5. Test circuit for opamps is a unity-gain follower configuration, with load resistor RL as specified.
+15V
R5
50K SIGNAL IN -15V
VCA SYM C2 R4
300K 47pF
C1
R1
20K0 1%
47uF
R3
51
R2
20K0 1% SIGNAL OUT
+15V
C7
OA1
IN EC-
SYM
OA3
VCA
OUT EC+
100n
+
VCC VEE
+
C8 C3
47uF 100n -15V
THAT4301
IN It
RMS
+
OUT Ct GND
OA2
R6
10K0 1%
-
C6
22uF
R7
2M00 1%
C4
10uF
RMS OUT Ec-
-15V
Figure 2. VCA and RMS detector test circuit
THAT Corporation; 45 Sumner Street; Milford, Massachusetts 01757-1656; USA Tel: +1 (508) 478-9200; Fax: +1 (508) 478-0990; Web: www.thatcorp.com
THAT 4301 Dynamics Processor IC
Page 5
Pin Name RMS In IT (ITime) No Connection RMS Out CT (CTime) OA2 -In OA2 Out OA2 +In GND VEE VCC OA3 Out VCA Out SYM EC+
DIP P in 1 2 3 4 5 6 7 8 9 10 11 12 13 14 15
SO Pin 3 4 5 6 7 9 10 11 12 13 18 19 20 22 23
Pin Name ECVCA In OA1 Out OA1 -In OA1 +In No Connection No Connection No Connection No Connection No Connection No Connection No Connection No Connection No Connection No Connection
DIP P in 16 17 18 19 20
SO Pin 24 25 26 27 28 1 2 8 14 15 16 17 21 29 30
Table 1. Pin Connections
E 1 F B A G
Ordering Information Plastic DIP Plastic Surface Mount 4301P 4301S 4301PA Inquire
0-10
H C D J K N L M P I
B 1
C I
O
D F A
E H G
J
0-15
ITEM A B C D E F G H I J K L M N O P
MILLIMETERS 24.8 Max. 24.2 +/-0.2 6.4 +/-0.2 7.62 +/-0.25 2.54 +/-0.15 0.46 +0.15 -0.1 1.0 +/-0.15 1.5 Typ. 0.98 Typ. 1.5 1.75 3.25 +/-0.15 4.7 Max. 0.51 Min. 2.8 Min. 0.25 +0.15 -0.05
INCHES 0.98 Max 0.95 +/-0.008 0.25 +/-0.008 0.30 +/-0.01 0.10 +/-0.006 0.02 +0.006 -0.004 0.04 +/-0.006 0.06 Typ. 0.04 Typ. 0.06 0.07 0.13 +/-0.006 0.19 Max. 0.02 Min. 0.11 Min. 0.01 +0.006 -0.002
ITEM A B C D E F G H I J
MILLIMETERS 15.4 +/- 0.3 7.5 +/- 0.2 10.3 +/- 0.4 0.4 + 0.1 - 0.05 1.0 Typ. 0.85 MAX. 2.3 +/- 0.15 0.15 +/- 0.1 0.8 0.2 + 0.1 - 0.05
INCHES 0.60 +/- 0.012 0.29 +/- 0.008 0.41 +/- 0.016 0.002 +0.004 -0.002 0.039 Typ. 0.033 Max. 0.09 +/- 0.006 0.006 +/- 0.004 0.031 0.008 +0.004 -0.002
Figure 3. Plastic dual in-line package outline
Figure 4. Plastic surface-mount package outline
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Page 6
Rev. 04/10/02
Representative Data
Figure 5. VCA Gain vs. Control Voltage (Ec-) at 25C
Figure 6. VCA 1kHz THD+Noise vs. Input, -15 dB Gain
Figure 7. VCA 1kHz THD+Noise vs. Input, +15 dB Gain
Figure 8. VCA 1kHz THD+Noise vs. Input, 0 dB Gain
Figure 9. VCA THD vs. Frequency, 0 dB Gain, 1Vrms Input
Figure 10. RMS Output vs. Input Level, 1 kHz & 10 kHz
Figure 11. Departure from Ideal Detector Law vs. Level
Figure 12. Detector Output vs. Frequency at Various Levels
THAT Corporation; 45 Sumner Street; Milford, Massachusetts 01757-1656; USA Tel: +1 (508) 478-9200; Fax: +1 (508) 478-0990; Web: www.thatcorp.com
THAT 4301 Dynamics Processor IC
Page 7
Theory of Operation
THAT 4301 Dynamics Processor combines THAT Corporation's proven Voltage-Controlled Amplifier (VCA) and RMS-Level Detector designs with three general-purpose opamps to produce an Analog Engine useful in a variety of dynamics processor applications. For details of the theory of operation of the VCA and RMS-Detector building blocks, the interested reader is referred to THAT Corporation's data sheets on the 2150 Series VCAs and the 2252 RMS-Level Detector. Theory of the interconnection of exponentially-controlled VCAs and log-responding level detectors is covered in THAT Corporation's application note AN101, The Mathematics of Log-Based Dynamic Processors. 2), which is adjusted for minimum signal distortion at unity (0 dB) gain. The VCA may be controlled via EC-, as shown in Figure 2, or via the combination of EC+ and SYM. This connection is illustrated in Figure 13. Note that this figure shows only that portion of the circuitry needed to drive the positive VCA control port; circuitry associated with OA1, OA2 and the RMS detector has been omitted.
R5 Positive Control In C1 Signal In 47uF 20K0 1% R3 51 SYM OUT VCA EC+ 50K VCA SYM C2 47pF R1 R4 300K R2 20K0 1% Signal Out
The VCA -- in Brief
THAT 4301 VCA is based on THAT Corporation's highly successful complementary log-antilog gain cell topology, as used in THAT 2150-Series IC VCAs, and the modular 202 Series VCAs. THAT 4301 is integrated using a fully complementary, BiFET process. The combination of FETs with high-quality, complementary bipolar transistors (NPNs and PNPs) allows additional flexibility in the design of the VCA over previous efforts. Input signals are currents to the VCA IN pin. This pin is a virtual ground, so in normal operation an input voltage is converted to input current via an appropriately sized resistor (R1 in Figure 2, Page 4). Because dc offsets present at the input pin and any dc offset in preceeding stages will be modulated by gain changes (thereby becoming audible as thumps), the input pin is normally ac-coupled (C1 in Figure 2). The VCA output signal is also a current, inverted with respect to the input current. In normal operation, the output current is converted to a voltage via inverter OA3, where the ratio of the conversion is determined by the feedback resistor (R2, Figure 2) connected between OA3`s output and its inverting input. The signal path through the VCA and OA3 is noninverting. The gain of the VCA is controlled by the voltage applied to EC-, EC+, and SYM. Gain (in decibels) is proportional to EC+ - EC-, provided EC+ and SYM are at essentially the same voltage (see below). The constant of proportionality is -6.5 mV/dB for the voltage at EC-, and 6.5 mV/dB for the voltage at EC+ and SYM. As mentioned, for proper operation, the same voltage must be applied to EC+ and SYM, except for a small (2.5 mV) dc bias applied between these pins. This bias voltage adjusts for internal mismatches in the VCA gain cell which would otherwise cause small differences between the gain of positive and negative half-cycles of the signal. The voltage is usually applied via an external trim potentiometer (R5 in Figure
OA1 + VCC VEE IN It
IN EC-
OA3 +
THAT4301
RMS
OUT Ct GND + OA2 -
Figure 13. Driving the VCA via the Positive Control Port
While the 4301's VCA circuitry is very similar to that of the THAT 2150 Series VCAs, there are several important differences, as follows: 1) Supply current for the VCA is fixed internally. Approximately 2mA is available for the sum of input and output signal currents. (This is also the case in a 2150 Series VCA when biased as recommended.) 2) The signal current output of the VCA is internally connected to the inverting input of an on-chip opamp. In order to provide external feedback around this opamp, this node is brought out to a pin. 3) The control-voltage constant is approximately 6.5 mV/dB, due primarily to the higher internal operating temperature of the 4301 compared to that of the 2150 Series. 4) The input stage of the 4301 VCA uses integrated P-channel FETs rather than a bias-current corrected bipolar differential amplifier. Input bias currents have therefore been reduced.
The RMS Detector -- in Brief
The 4301's detector computes rms level by rectifying input current signals, converting the rectified current to a logarithmic voltage, and applying that voltage to a log-domain filter. The output signal is a dc voltage proportional to the decibel-level of the rms
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Page 8
Rev. 04/10/02
value of the input signal current. Some ac component (at twice the input frequency) remains superimposed on the dc output. The ac signal is attenuated by a log-domain filter, which constitutes a single-pole rolloff with cutoff determined by an external capacitor and a programmable dc current. As in the VCA, input signals are currents to the RMS IN pin. This input is a virtual ground, so a resistor (R6 in Figure 2) is normally used to convert input voltages to the desired current. The level detector is capable of accurately resolving signals well below 10 mV (with a 10 kW input resistor). However, if the detector is to accurately track such low-level signals, ac coupling is normally required. The log-domain filter cutoff frequency is usually placed well below the frequency range of interest. For an audio-band detector, a typical value would be 5 Hz, or a 32 ms time constant (t). The filter's time constant is determined by an external capacitor attached to the CT pin, and an internal current source (ICT) connected to CT. The current source is programmed via the IT pin: current in IT is mirrored to ICT with a gain of approximately 1.1. The resulting time constant t is approximately equal to 0.026 CT/IT. Note that, as a result of the mathematics of RMS detection, the attack and release time constants are fixed in their relationship to each other. The dc output of the detector is scaled with the same constant of proportionality as the VCA gain control: 6.5 mV/dB. The detector's 0 dB reference (Iin0, the input current which causes 0 V output), is determined by IT as follows: Iin0= 9.6 mA It The detector output stage is capable of sinking or sourcing 100 mA. Differences between the 4301's RMS-Level Detector circuitry and that of the THAT 2252 RMS Detector are as follows: 1) The rectifier in the 4301 RMS Detector is internally balanced by design, and cannot be balanced via an external control. The 4301 will typically balance positive and negative halves of the input signal within 1.5%, but in extreme cases the mismatch may reach 15%. However, a 15% mismatch will not significantly increase ripple-induced distortion in dynamics processors over that caused by signal
ripple alone. 2) The time constant of the 4301's RMS detector is determined by the combination of an external capacitor (connected to the CT pin) and an internal, programmable current source. The current source is equal to 1.1 IT. Normally, a resistor is not connected directly to the CT pin on the 4301. 3) The 0 dB reference point, or level match, is not adjustable via an external current source. However, as in the 2252, the level match is affected by the timing current, which, in this case, is drawn from the IT pin and mirrored internally to CT. 4) The input stage of the 4301 RMS detector uses integrated P-channel FETs rather than a bias-current corrected bipolar differential amplifier. Input bias currents are therefore negligible, improving performance at low signal levels.
The Opamps -- in Brief
The three opamps in the 4301 are intended for general purpose applications. All are 5 MHz opamps with slew rates of approximately 2V/ms. All use bipolar PNP input stages. However, the design of each is optimized for its expected use. Therefore, to get the most out of the 4301, it is useful to know the major differences among these opamps. OA3, being internally connected to the output of the VCA, is intended for current-to-voltage conversion. Its input noise performance, at 7.5nV Hz, complements that of the VCA, adding negligible noise at unity gain. Its output section is capable of driving a 2 kW load to within 2V of the power supply rails, making it possible to use this opamp directly as the output stage in single-ended designs. OA1 is the quietest opamp of the three. Its input noise voltage, at 6.5nV Hz, makes it the opamp of choice for input stages. Note that its output drive capability is limited (in order to reduce the chip's power dissipation) to approximately 3 mA. It is comfortable driving loads of 5 kW or more to within 1V of the power supply rails. OA2 is intended primarily as a control-voltage processor. Its input noise parallels that of OA3, and its output drive capability parallels that of OA1.
THAT Corporation; 45 Sumner Street; Milford, Massachusetts 01757-1656; USA Tel: +1 (508) 478-9200; Fax: +1 (508) 478-0990; Web: www.thatcorp.com
THAT 4301 Dynamics Processor IC
Page 9
Applications
The circuit of Figure 14, Page 9, shows a typical application for THAT 4301. This simple compressor/limiter design features adjustable hard-knee threshold, compression ratio, and static gain1. The applications discussion in this data sheet will center on this circuit for the purpose of illustrating important design issues. However, it is posslble to configure many other types of dynamics processors with THAT 4301. Hopefully, the following discussion will imply some of these possibilities. verting stage. If, for some reason, more than 0 dB gain is required when the VCA is set to unity, then the resistors may be skewed to provide it. Note that the feedback capacitor (C2) is required for stability. The VCA output has approximately 45 pf of capacitance to ground, which must be neutralized via the 47 pf feedback capacitor across R2. The VCA gain is controlled via the EC- terminal, whereby gain will be proportional to the negative of the voltage at EC-. The EC+ terminal is grounded, and the SYM terminal is returned nearly to ground via a small resistor (R3, 51 W). The VCA SYM trim (R5, 50 kW) allows a small voltage to be applied to the SYM terminal via R4 (300 kW). This voltage adjusts for small mismatches within the VCA gain cell, thereby reducing even-order distortion products. To adjust the trim, apply to the input a middle-level, middle-frequency signal (1 kHz at 1 V is a good
Signal Path
As mentioned in the section on theory, the VCA input pin is a virtual ground with negative feedback provided internally. An input resistor (R1, 20kW) is required to convert the ac input voltage to a current within the linear range of the 4301. (Peak VCA input currents should be kept under 1 mA for best distortion performance.) The coupling capacitor (C1, 47 mf) is strongly recommended to block dc current from preceeding stages (and from offset voltage at the input of the VCA). Any dc current C1 into the VCA will be 47uF modulated by varying +15 gain in the VCA, showing THRESHOLD CCW up in the output as R11 R12 "thumps". Note that C1, 383K 1% 10K R10 in conjunction with R1, CW 2M00 1% will set the low fre-15 R8 quency limit of the cir4k99 1% cuit. IN The VCA output is connected to OA3, configured as an inverting current-to-voltage converter. OA3`s feedback components (R2, 20 kW, and C2, 47 pf) determine the constant of current-to-voltage conversion. The simplest way to deal with this is to recognize that when the VCA is set for unity (0 dB) gain, the input to output voltage gain is simply R2/R1, just as in the case of a single inC7 C8 C3 47uF +15 100n 100n -15 R6 10K0 1% C6 R14 1K43 1% 22uF
VCA SYM R1 20K0 1% R9 10K0 1% CR2 CR1 -15 R4 300K R3 51 R2 20K0 1% OA1 + VCC VEE IN IT SYM OUT EC+ OA3 + OUT C2 47pF R5 50K +15
IN EC-
VCA
THAT4301
RMS OUT
CT GND + OA2 R16
R7 2M00 1% -15
4k99 1% C4 10uF C5 100N R17 R15 590K 1% +15 GAIN
CW
COMPRESSION R13 10K
CW
R18 10K
CCW
CCW
10K0 1%
-15
Figure 14. Typical Compressor/Limiter Application Circuit
1. More information on this compressor design, along with suggestions for converting it to soft-knee operation, is given in AN100, Basic Compressor Limiter Design. The designs in AN100 are based on THAT Corporation's 2150-Series VCAs and 2252 RMS Detector, but are readily adaptable to the 4301 with only minor modifications. In fact, the circuit presented here is functionally identical to the hard-knee circuit published in AN100.
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Page 10
Rev. 04/10/02
choice with this circuit) and observe THD at the signal output. Set the trim for minimum THD.
RMS-Level Detector
The RMS detector's input is similar to that of the VCA. An input resistor (R6, 10 kW) converts the ac input voltage to a current within the linear range of the 4301. (Peak detector input currents should be kept under 1 mA for best linearity.) The coupling capacitor (C3, 47 mf) is recommended to block dc current from preceeding stages (and from offset voltage at the input of the detector). Any dc current into the detector will limit the low-level resolution of the detector, and will upset the rectifier balance at low levels. Note that, as with the VCA input circuitry, C3 in conjunction with R6 will set the lower frequency limit of the detector. The time response of the RMS detector is determined by the capacitor attached to CT (C4, 10 mf) and the size of the current in pin IT (determined by R7, 2 MW and the negative power supply, -15V). Since the voltage at IT is approximately 0 V, the circuit of Figure 14 produces 7.5 mA in IT. The current in IT is mirrored with a gain of 1.1 to the CT pin, where it is available to discharge the timing capacitor (C4). The combination produces a log filter with time constant equal to approximately 0.026 CT/IT (~35 ms in the circuit shown). The waveform at CT will follow the logged (decibel) value of the input signal envelope, plus a dc offset of about 1.3 V (2 VBE). This allows a polarized capacitor to be used for the timing capacitor, usually an electrolytic. The capacitor used should be a low-leakage type in order not to add significantly to the timing current. The output stage of the RMS detector serves to buffer the voltage at CT and remove the 1.3 V dc offset, resulting in an output centered around 0 V for input signals of about 85 mV. The output voltage increases 6.5 mV for every 1 dB increase in input signal level. This relationship holds over more than a 60 dB range in input currents.
loop such that the junction of R9 and CR2 (the output of the threshold detector) sits at -(R9/R8) RMSOUT. For the circuit of Figure 14, this is -2 RMSOUT. Negative signals from the RMS detector drive the output of OA1 positive, reverse biasing CR2 and forward biasing CR1. In this case, the junction of R9 and CR2 rests at 0 V, and no signal level informaion is passed to the threshold detector's output. In order to vary the threshold, R12, the THRESHOLD control, is provided. Via R11 (383 kW), R12 adds up to 39.2 mA of current to OA1`s summing junction, requiring the same amount of opposite-polarity current from the RMS detector output to counterbalance it. At 4.99 kW, the voltage across R8 required to produce a counterbalancing current is 195 mV, which represents a 30 dB change in RMS detector input level. Since the RMS detector's 0 dB reference level is 85 mV, the center of the THRESHOLD pot's range would be 85 mV, were it not for R10 (2 MW), which provides an offset. R10 adds an extra -7.5 ma to OA1`s summing junction, which would be counterbalanced by 37.4 mV at the detector output. This corresponds to 5.8 dB, offsetting the THRESHOLD center by this much to 165 mV, or approximately -16 dBV. The output of the threshold detector represents the signal level above the determined threshold, at a constant of about 13 mV/dB (from [R9/R8] 6.5 mV/dB). This signal is passed on to the COMPRESSION control (R13), which variably attenuates the signal passed on to OA2. Note that the gain of OA2, from the wiper of the COMPRESSION control to OA2`s output, is R16/R15 (0.5), precisely the inverse of the gain of OA1. Therefore, the COMPRESSION control lets the user vary the above-threshold gain between the RMS detector output and the output of OA1 from zero to a maximum of unity. The gain control constant of the VCA, 6.5 mV/dB, is exactly equal to the output scaling constant of the RMS detector. Therefore, at maximum COMPRESSION, above threshold, every dB increase in input signal level causes a 6.5 mV increase in the output of OA2, which in turn causes a 1 dB decrease in the VCA gain. With this setting, the output will not increase despite large increases in input level above threshold. This is infinite compression. For intermediate settings of COMPRESSION, a 1 dB increase in input signal level will cause less than a 1 dB decrease in gain, thereby varying the compression ratio. The resistor R14 is included to alter the taper of the COMPRESSION pot to better suit common use. If a linear taper pot is used for R13, the compression ratio will be 1:2 at the middle of the rotation. However, 1:2 compression in an above-threshold compressor is not very strong processing, so 1:4 is often preferred at the midpoint. R14 warps the taper of R13 so that 1:4 compression occurs at approximately the midpoint of R13`s rotation.
Control Path
A compressor/limiter is intended to reduce its gain as signals rise above a threshold. The output of the RMS detector represents the input signal level over a wide range of levels, but compression only occurs when the level is above the threshold. OA1 is configured as a variable threshold detector to block envelope information for low-level signals, passing only information for signals above threshold. OA1 is an inverting stage with gain of 2 above threshold and 0 below threshold. Neglecting the action of the THRESHOLD control (R12) and its associated resistors (R11 and R10), positive signals from the RMS detector output drive the output of OA1 negative. This forward biases CR2, closing the feedback
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THAT 4301 Dynamics Processor IC
Page 11
The GAIN control (R18) is used to provide static gain or attenuation in the signal path. This control adds up to 130 mV offset to the output of OA2 (from -V+ R 16 R to -V- R 16 R ), which is approximately
17 17
20 dB change in gain of the VCA. C5 is used to attenuate the noise of OA2, OA1 and the resistors R8 through R16 used in the control path. All these active and passive components produce noise which is passed on to the control port of the VCA, causing modulation of the signal. By itself, the 4301 VCA produces very little noise modulation, and its performance can be significantly degraded by the use of noisy components in the control voltage path.
may be varied from 1:1 (no compression) to :1. And, static gain may be added up to 20 dB. Audio performance is excellent, with THD running below 0.05% at middle frequencies even with 10 dB of compression, and an input dynamic range of over 115 dB. Perhaps most important, this example design only scratches the surface of the large body of applications circuits which may be constructed with THAT 4301. The combination of an accurate, wide-dynamic-range, log-responding level detector with a high-quality, exponentially-responding VCA produces a versatile and powerful analog engine. The opamps provided in the 4301 enable the designer to configure these building blocks with few external components to construct gates, expanders, de-essers, noise reduction systems and the like. For further information, samples and pricing, please contact us at the address below.
Overall Result
The resulting compressor circuit provides hard-knee compression above threshold with three essential user-adjustable controls. The threshold of compression may be varied over a 30 dB range from about -46 dBV to +14 dBV. The compression ratio
THAT Corporation; 45 Sumner Street; Milford, Massachusetts 01757-1656; USA Tel: +1 (508) 478-9200; Fax: +1 (508) 478-0990; Web: www.thatcorp.com
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Rev. 04/10/02
Notes
THAT Corporation; 45 Sumner Street; Milford, Massachusetts 01757-1656; USA Tel: +1 (508) 478-9200; Fax: +1 (508) 478-0990; Web: www.thatcorp.com

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